| Home |
| Search |
| Today's Posts |
|
#1
Posted to rec.audio.tubes
|
|||
|
|||
12AX7 grid current
I am having a strange problem with what appears to be distortion due to grid current in a 12AX7.
It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? Cheers Ian |
|
#2
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? I could spend a while calculating the drop in grid resistance and the rise in grid voltage due to signal, and compare the two, but maybe you've already done that. What's the grid resistance or current *supposed* to be, and how should it change with voltage? Do all the grids of all the valves in the world behave in broadly the same way, regardless of their purpose? At low anode voltage does grid current rise steeply? Are you using a typical operating point and load, or have you made it up yourself? 120k seems very low for an ECC83. Do you need the 100k? If so, could you use a lower fixed resistance multiplied by feedback, and could that arrangement alleviate the grid current problem? The only mention I found of grid current in datasheets is on page two in this: http://www.tubezone.net/pdf/12ax7ecc83.pdf Where it appears to note (*) that the distortion shown in the table is due to grid current. Another thing, in passing, is that your test circuit had me musing on the difference between a grid stopper and a source resistance. Calculations would be easier if you swapped that 100k to the other side of the 1meg leak, maybe. Ian |
|
#3
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 16, 5:34*am, Ian Bell wrote:
I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer *is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? Cheers Ian In some old AM radio designs where the first audio tube was a high gain triode like a 12AX7 the method of biasing was often grid leak bias with a 10M Rg and no Rk was used. There is a slight current flow along the 10M giving a bias of about -1V and this bias current is about proportional to Ia so the more Ia, the more -bias and so the triode then self biases OK. The distortion generated was ignored and was much less than from other causes in most old radios. The onset of grid current in a 12AX7 is gradual with the actual bias voltage value, and Ig begins at some Eg1 well below 0V, and not only when Eg1 goes positive with respect to Ek. 12AX7 are rather a beancounter's favourite tube because you get a load of gain. But maybe you'll find that the finite input resistance which you are discovering exists also appears with other triodes. The fact the distortion dissapears when you short out the series 100k is because your source impedance from sig gene falls to maybe600 ohms and the grid input resistance is virtually completely shunted, and the Ia is then controlled without the non linear grid input resistance interfering. To get the best with 12AX7 performance one should always try to get Ea rather high for a given amount of Ia which should be more than 0.6mA, so that a decent negative bias voltage to the grid is needed. Whatever is driving the grid of the 'AX7 should have low source resisance. Now have you tried biasing the 'AX7 grid with say 47k? This is a typical value for a phono amp, and could also be good for the input of a power amp although you'll commonly see 470k used for power amps and line stages. If you have a source select switch or a switched volume control in the grid circuit of some preamp or power amp input tubes you can sometimes hear a click each time the source is changed or the volume is adjusted and its the altering level of grid Vdc. It only takes a jerk of a few mV to make a click sound. To avoid such nonsense, one might use say 100k grid bias R with say 0.22uF input blocking cap with source select and volume control in front of the cap and not able to very slightly adjust whatever slight Vdc may exist at the grid. And if you want lowest possible noise generated by the grid biasing resistance you need to keep the R value low as possible but without being too low a load for the driving stage. If this is a CD player with Rout less than 600 ohms you could have Rg = 10k and maybe no need for any dc blocker cap. As tubes age, the grid can act strangely and its slight tendency to be always slightly more negative than the bias supply, or say 1V ess than 0V where Rg = 10M, becomes less of a tendency, and the bias control becomes lost and grid slowly becomes positive and Ia then is turned on more so that older tubes can have a positive grid with very low Ea and high Ia as a result. This happens in old radios and nobody notices until Ea has declined to about +20V only, and distortion is beginning to get audible at a percent or more. Your 12AX7 operating conditions seem about just right, and I guess changing samples of 12AX7 should not change the phenomena you see - although some well worn 'AX7 samples will probably measure more THD than newer ones with a harder vacuum. Things could be worse - if you used a small BJT with bypassed emitter resistance - the THD will be horrendous, but with a much bigger difference between having a 100k in series with source and base and having no series R. Patrick Turner. |
|
#4
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Iveson wrote:
Ian Bell wrote: I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? I could spend a while calculating the drop in grid resistance and the rise in grid voltage due to signal, and compare the two, but maybe you've already done that. Yes, we should be well away from where grid current normally starts. What's the grid resistance or current *supposed* to be, and how should it change with voltage? Do all the grids of all the valves in the world behave in broadly the same way, regardless of their purpose? At low anode voltage does grid current rise steeply? Are you using a typical operating point and load, or have you made it up yourself? 120k seems very low for an ECC83. 120k is about right. The data sheet you quote below gives figures for 47k, 100k and 200K so 120K is not unuasual. Do you need the 100k? If so, could you use a lower fixed resistance multiplied by feedback, and could that arrangement alleviate the grid current problem? The 100K is there as the input arm of an eventual NFB network. The only mention I found of grid current in datasheets is on page two in this: http://www.tubezone.net/pdf/12ax7ecc83.pdf Where it appears to note (*) that the distortion shown in the table is due to grid current. That's a nicely copmplete datasheet. Thanks. Another thing, in passing, is that your test circuit had me musing on the difference between a grid stopper and a source resistance. Calculations would be easier if you swapped that 100k to the other side of the 1meg leak, maybe. Ian Cheers ian |
|
#5
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 16, 5:34 am, Ian wrote: I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? Cheers Ian In some old AM radio designs where the first audio tube was a high gain triode like a 12AX7 the method of biasing was often grid leak bias with a 10M Rg and no Rk was used. There is a slight current flow along the 10M giving a bias of about -1V and this bias current is about proportional to Ia so the more Ia, the more -bias and so the triode then self biases OK. The distortion generated was ignored and was much less than from other causes in most old radios. The onset of grid current in a 12AX7 is gradual with the actual bias voltage value, and Ig begins at some Eg1 well below 0V, and not only when Eg1 goes positive with respect to Ek. 12AX7 are rather a beancounter's favourite tube because you get a load of gain. But maybe you'll find that the finite input resistance which you are discovering exists also appears with other triodes. The fact the distortion dissapears when you short out the series 100k is because your source impedance from sig gene falls to maybe600 ohms and the grid input resistance is virtually completely shunted, and the Ia is then controlled without the non linear grid input resistance interfering. Yes, I have come across the same effect in my work with mu followers. I was just surprised to see it acting so soon. generally you can allow the signal to take the grid up to -1V before worrying about grid current bu that seems not to bee the case with the 12AX7 - today I fed it with a mere 25mV and it produced 1% distortion. There's something funny going on here. To get the best with 12AX7 performance one should always try to get Ea rather high for a given amount of Ia which should be more than 0.6mA, so that a decent negative bias voltage to the grid is needed. Looking at the curves that is going to be quite hard. To get -2V bias at 200V then Ia needs to be way below 0.6mA. Whatever is driving the grid of the 'AX7 should have low source resisance. That is n ot an option in this case as it will be a 100K volume control so even if that is fed from zero ohms the worst case source R would be 25K. Now have you tried biasing the 'AX7 grid with say 47k? This is a typical value for a phono amp, and could also be good for the input of a power amp although you'll commonly see 470k used for power amps and line stages. I am using 1Meg at present - ah maybe that's what is causing the trouble - I made it that big so as not to disturb the NFB that will be added later - the 100K would be the input arm of the NFB - I'll try smaller grid R and see if that helps - good idea - thanks Patrick. If you have a source select switch or a switched volume control in the grid circuit of some preamp or power amp input tubes you can sometimes hear a click each time the source is changed or the volume is adjusted and its the altering level of grid Vdc. It only takes a jerk of a few mV to make a click sound. To avoid such nonsense, one might use say 100k grid bias R with say 0.22uF input blocking cap with source select and volume control in front of the cap and not able to very slightly adjust whatever slight Vdc may exist at the grid. And if you want lowest possible noise generated by the grid biasing resistance you need to keep the R value low as possible but without being too low a load for the driving stage. If this is a CD player with Rout less than 600 ohms you could have Rg = 10k and maybe no need for any dc blocker cap. It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. As tubes age, the grid can act strangely and its slight tendency to be always slightly more negative than the bias supply, or say 1V ess than 0V where Rg = 10M, becomes less of a tendency, and the bias control becomes lost and grid slowly becomes positive and Ia then is turned on more so that older tubes can have a positive grid with very low Ea and high Ia as a result. This happens in old radios and nobody notices until Ea has declined to about +20V only, and distortion is beginning to get audible at a percent or more. Don't think that is a problem as I am using new tubes right now. Your 12AX7 operating conditions seem about just right, and I guess changing samples of 12AX7 should not change the phenomena you see - although some well worn 'AX7 samples will probably measure more THD than newer ones with a harder vacuum. As I said both ones I tried are new - both by EH. Things could be worse - if you used a small BJT with bypassed emitter resistance - the THD will be horrendous, but with a much bigger difference between having a 100k in series with source and base and having no series R. LOL Cheers Ian Patrick Turner. |
|
#6
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote in message
I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? I could spend a while calculating the drop in grid resistance and the rise in grid voltage due to signal, and compare the two, but maybe you've already done that. Yes, we should be well away from where grid current normally starts. What's the grid resistance or current *supposed* to be, and how should it change with voltage? Do all the grids of all the valves in the world behave in broadly the same way, regardless of their purpose? At low anode voltage does grid current rise steeply? Are you using a typical operating point and load, or have you made it up yourself? 120k seems very low for an ECC83. 120k is about right. The data sheet you quote below gives figures for 47k, 100k and 200K so 120K is not unuasual. Hmm. The EEC83 has been used for just about every purpose, for which there are many optimal conditions, I guess. Usually, for hi-fi, I would expect a triode to have a load of between two and three times its anode resistance. That would make 220k, 270k, or even higher. That in turn means a higher HT and low current. Looking at a couple of McIntoshes, I see they are radically different: http://www.pmillett.com/file_downloa...h/A116_sch.pdf (don't know where those voltages came from...) http://www.pmillett.com/file_downloa...sh/C20_sch.pdf which is more what I expected to see. A visit to http://tdsl.duncanamps.com/schematics.php gives plenty of examples of what kind of operating points are used for what purpose. Guitar amps seem to go for 100k. Eico hi-fi amps, using EEC83 come hell or high water, start with 220k then drop to around 100k for subsequent stages that need to drive a bit of current. You didn't comment on my suggestion that a higher Vak might help to reduce grid current. Works for screen grids. How about an active load? What impedance are you driving? Flipper's mention of capacitance might be worth following up. Does your effect depend on frequency? Is "non-linear capacitance" the same thing as "linear capacitance plus the same non-linear behaviour we were talking about in the first place"? In construction terms, what's the key difference between high and low gain valves? Is the grid closer to the anode? Unless you provide those electrons with somewhere more attractive to go, you'll have to sink them somewhere. Finally, did you say bootstrap? Won't that reduce the source resistance seen by the grid? And what's the impedance of your probe? Ian |
|
#7
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
snip for brevity....
Whatever is driving the grid of the 'AX7 should have low source resisance. That is n ot an option in this case as it will be a 100K volume control so even if that is fed from zero ohms the worst case source R would be 25K. Yes but in normal use the volume control would be at about the midday position or -20dB attenuation position. The Rout from a 100k pot at this position where the R between wiper and 0V is about 9k and the top part of the pot is 90k to a low Z source so Rout is 90k // 9k = 8 k approx which I suspect is low enough to shunt the apparent non linear input resistance which makes your distortion a bit too high. I've just been called out to do a house call and will continue a reply to the rest of your post later. Patrick Turner. Now have you tried biasing the 'AX7 grid with say 47k? This is a typical value for a phono amp, and could also be good for the input of a power amp although you'll commonly see 470k used for power amps and line stages. I am using 1Meg at present - ah maybe that's what is causing the trouble - I made it that big so as not to disturb the NFB that will be added later - the 100K would be the input arm of the NFB - I'll try smaller grid R and see if that helps - good idea - thanks Patrick. * If you have a source select switch or a switched volume control in the grid circuit of some preamp or power amp input tubes you can sometimes hear a click each time the source is changed or the volume is adjusted and its the altering level of grid Vdc. It only takes a jerk of a few mV to make a click sound. To avoid such nonsense, one might use say 100k grid bias R with say 0.22uF input blocking cap with source select and volume control in front of the cap and not able to very slightly adjust whatever slight Vdc may exist at the grid. And if you want lowest possible noise generated by the grid biasing resistance you need to keep the R value low as possible but without being too low a load for the driving stage. If this is a CD player with Rout less than 600 ohms you could have Rg = 10k and maybe no need for any dc blocker cap. It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. As tubes age, the grid can act strangely and its slight tendency to be always slightly more negative than the bias supply, or say 1V ess than 0V where Rg = 10M, becomes less of a tendency, and the bias control becomes lost and grid slowly becomes positive and Ia then is turned on more so that older tubes can have a positive grid with very low Ea and high Ia as a result. This happens in old radios and nobody notices until Ea has declined to about +20V only, and distortion is beginning to get audible at a percent or more. Don't think that is a problem as I am using new tubes right now. Your 12AX7 operating conditions seem about just right, and I guess changing samples of 12AX7 should not change the phenomena you see - although some well worn 'AX7 samples will probably measure more THD than newer ones with a harder vacuum. As I said both ones I tried are new - both by EH. Things could be worse - if you used a small BJT with bypassed emitter resistance - the THD will be horrendous, but with a much bigger difference between having a 100k in series with source and base and having no series R. LOL Cheers Ian Patrick Turner.- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#8
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
flipper wrote:
On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: snip Yes, I have come across the same effect in my work with mu followers. I was just surprised to see it acting so soon. generally you can allow the signal to take the grid up to -1V before worrying about grid current bu that seems not to bee the case with the 12AX7 - today I fed it with a mere 25mV and it produced 1% distortion. There's something funny going on here. You are chasing phantoms as it isn't 'positive grid drive'. It's called a "grid leak" resistor for a reason: there's grid (leak) current. (This, btw, is what determines the 'recommended maximum' grid leak resistor.) It's small but there nonetheless and, being roughly proportional to plate current, varies with signal as well. As for the size of it, RDH4 recommends that, for cathode bias 'high mu' triodes, the grid leak be no larger than 3x the (DC) plate load, and no larger than 4x the preceding stage's plate load, but that's a bias shift consideration and larger values were common practice. As for distortion, what you've got on the grid is a summing junction of your input and the 'grid leak' signal plus, of course, inter electrode capacitances so any measurement there will, of necessity, be 'distorted' relative to your source. Lower the source impedance and you increase it's proportion of the sum so the relative distortion decreases but it will never be '0' unless you turn off the tube (all of which you've observed). Why are you trying to measure distortion on the grid anyway? Only because I traced back to there starting from the ouptut. Cheers Ian |
|
#9
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Iveson wrote:
Ian Bell wrote in message I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? I could spend a while calculating the drop in grid resistance and the rise in grid voltage due to signal, and compare the two, but maybe you've already done that. Yes, we should be well away from where grid current normally starts. What's the grid resistance or current *supposed* to be, and how should it change with voltage? Do all the grids of all the valves in the world behave in broadly the same way, regardless of their purpose? At low anode voltage does grid current rise steeply? Are you using a typical operating point and load, or have you made it up yourself? 120k seems very low for an ECC83. 120k is about right. The data sheet you quote below gives figures for 47k, 100k and 200K so 120K is not unuasual. Hmm. The EEC83 has been used for just about every purpose, for which there are many optimal conditions, I guess. Usually, for hi-fi, I would expect a triode to have a load of between two and three times its anode resistance. That would make 220k, 270k, or even higher. That in turn means a higher HT and low current. To avoid higher HT I am bootstrapping the anode load from the cathode of the white follower that it drives. Looking at a couple of McIntoshes, I see they are radically different: http://www.pmillett.com/file_downloa...h/A116_sch.pdf (don't know where those voltages came from...) http://www.pmillett.com/file_downloa...sh/C20_sch.pdf which is more what I expected to see. Yes the latter one has the high anode resistors and low grid resistor. The former is a classic way of avoiding one of the zeros in the loop gain that give so much problem for NFB in tubes. The upshot is you have to have a dc path through the NFB which means you end uo with the first cathode 15V or so above ground. A visit to http://tdsl.duncanamps.com/schematics.php gives plenty of examples of what kind of operating points are used for what purpose. Guitar amps seem to go for 100k. Eico hi-fi amps, using EEC83 come hell or high water, start with 220k then drop to around 100k for subsequent stages that need to drive a bit of current. You didn't comment on my suggestion that a higher Vak might help to reduce grid current. Works for screen grids. How about an active load? What impedance are you driving? Vak is already higher than I want it to be. At present it is dc coupled to a white follower which really needs about 150V on the top grid. Flipper's mention of capacitance might be worth following up. Does your effect depend on frequency? Is "non-linear capacitance" the same thing as "linear capacitance plus the same non-linear behaviour we were talking about in the first place"? Not checked out frequency dependence yet. In construction terms, what's the key difference between high and low gain valves? Is the grid closer to the anode? Unless you provide those electrons with somewhere more attractive to go, you'll have to sink them somewhere. Finally, did you say bootstrap? Won't that reduce the source resistance seen by the grid? Indeed not, it increases the sac value of the anode load. And what's the impedance of your probe? The usual 1Meg plus a few pF Cheers Ian Ian |
|
#10
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
flipper wrote:
On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: snip Yes, I have come across the same effect in my work with mu followers. I was just surprised to see it acting so soon. generally you can allow the signal to take the grid up to -1V before worrying about grid current bu that seems not to bee the case with the 12AX7 - today I fed it with a mere 25mV and it produced 1% distortion. There's something funny going on here. You are chasing phantoms as it isn't 'positive grid drive'. It's called a "grid leak" resistor for a reason: there's grid (leak) current. (This, btw, is what determines the 'recommended maximum' grid leak resistor.) It's small but there nonetheless and, being roughly proportional to plate current, varies with signal as well. As for the size of it, RDH4 recommends that, for cathode bias 'high mu' triodes, the grid leak be no larger than 3x the (DC) plate load, and no larger than 4x the preceding stage's plate load, but that's a bias shift consideration and larger values were common practice. As for distortion, what you've got on the grid is a summing junction of your input and the 'grid leak' signal plus, of course, inter electrode capacitances so any measurement there will, of necessity, be 'distorted' relative to your source. Lower the source impedance and you increase it's proportion of the sum so the relative distortion decreases but it will never be '0' unless you turn off the tube (all of which you've observed). Why are you trying to measure distortion on the grid anyway? Only because I traced back to there starting from the ouptut. What distortion did you get on the plate and what were you expecting it to be? Exactly the same as at the grid - about 2H at -30dB or worse. If I short the input resistor 2H at the grid drops below -70dB and at the anode falls to -40dB which is about what I would expect. So I was expecting -40dB 2H at the anode and got a lot higher so traced back to the grid and found it was the same but that it wnet away with near zero source impedance. Cheers ian Cheers Ian |
|
#11
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 17, 6:58*am, Ian Bell wrote:
Patrick Turner wrote: On Sep 16, 5:34 am, Ian *wrote: snip a bit, and now reply to the rest of your post.... It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. That's interesting. I've never tested a White follower for its distortion, but Dn should be low. I would have maybe thought a 12BH7 a little weak for headphones and a direct connection to a 32 ohm load. Are you using an OPT? The load for any kind of follower should be still at least above 2 x Ra, and BH7 has Ra = about 5k per half, so RL should be 10k minimum for 1 triode section and therefore 5k for the pair of triodes. And 5k when driven with a White follower should give very low Dn. A pair of trioded EL84 make a nice white follower with 25mA for Ia and about Ea = about 200V per tube. Load can then be about 1k, capacitor coupled off the cathode-anode joint between the two triodes. I did use a White follower as an output buffer in a wien bridge oscillator I once made. This buffer had a pair of EL36 or 6CM5 and its contribution to Dn was negligible. But I later thought it was simpler and just as effective to make an oscillator buffer to be the driver of the positive and negative FB networks of the wien bridge oscillator and thus get a much higher possible top frequency which is now 250kHz. The trouble with wien bridge oscillators and HF is that the R&C networks for the FB become low impedance at F goes higher and this loads an output gain pentode adversely, especially if one wants about 14Vrms of output which is so easy at 1kHz, and RC networks which hardly loads the output gain pentode. My oscillator uses a 6AC7 directly coupled to the CF output buffer. The buffer has a dc load with a CC sink, so its output mainly only sees the cap coupled FB loops and the potentiometers which adjust the output to the outside world. I've been meaning to couple together a few of the identical 3 gang radio tuning caps I have to make a much better wien bridge oscillator which will have a higher top frequency and a much lower bottom frequency. I find the stability of operation of the wien bridge oscillator is best when using radio tuning caps rather than trying to use a dual gang potentiometer. But time is my enemy. Patrick Turner. As tubes age, the grid can act strangely and its slight tendency to be always slightly more negative than the bias supply, or say 1V ess than 0V where Rg = 10M, becomes less of a tendency, and the bias control becomes lost and grid slowly becomes positive and Ia then is turned on more so that older tubes can have a positive grid with very low Ea and high Ia as a result. This happens in old radios and nobody notices until Ea has declined to about +20V only, and distortion is beginning to get audible at a percent or more. Don't think that is a problem as I am using new tubes right now. Your 12AX7 operating conditions seem about just right, and I guess changing samples of 12AX7 should not change the phenomena you see - although some well worn 'AX7 samples will probably measure more THD than newer ones with a harder vacuum. As I said both ones I tried are new - both by EH. Things could be worse - if you used a small BJT with bypassed emitter resistance - the THD will be horrendous, but with a much bigger difference between having a 100k in series with source and base and having no series R. LOL Cheers Ian Patrick Turner.- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#12
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 18, 8:39*am, Ian Bell wrote:
flipper wrote: On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: Snip for brevity. Exactly the same as at the grid - about 2H at -30dB or worse. If I short the input resistor 2H at the grid drops below -70dB and at the anode falls to -40dB which is about what I would expect. So I was expecting -40dB 2H at the anode and got a lot higher so traced back to the grid and found it was the same but that it wnet away with near zero source impedance. Your variations of Dn with input series resistance for 12AX7 is interesting, and not many ppl would be wise to it. But what of other tubes? In all my tests for line level preamps using 12AU7, 6SN7, and my favourite, 6CG7, the Dn you mention as a result of high input source resistance never seemed to occur. What if you change from 12AX7 to 12AU7? The 'AU7 should work in your circuit with minor load&biasing changes. I recall measuring a 6CG7 line stage with 100k volume control before a CC gain tube using 47k anode load then direct coupled CF to output and at typical levels to drive a power amp at normal listening levels Dn was always less than 0.02%, or about -74dB, which I thought would always be inaudible, and completely overwhelmed by the Dn of the power amp and speakers. BTW, the amp with 6CG7 was tried with 4 different brands of 6CG7 and with music and with 4 of us listening for an afternoon. There was a considerable difference to the sound between excellent with Siemans NOS and not very good at all with Sovtek, but I thought an Aust made was right up there with the Siemans. The NOS Mullard was just polite sounding. all tubes would not have had very different Dn profiles. But I have never built a preamp with 100k of series input resistance. But then there is a another consideration. Many line stages are made using a gain tube which may or may not have a follower buffer to its outout. And there may well be a volume control at the input. and after the volume control the gain tube grid has a 100k series R to its grid and say a 560k taken to the buffer output. This is a simple shunt FB network and it gives closed loop gain of about 5x, and if the gain triode was a 12AX7 with open loop gain = 70x, then gain reduction with FB = 1/14, and any Dn is also reduced by the same margin UNLESS of course the series input resistance increases the Dn which would result in much less Dn reduction than a factor of 1/14. Baxandall tone control stages may typically have a CF input stage or low Rout gain stage driving the input to a shunt FB network around a tone control gain tube. The potentiometers used for the tone control are linear pots and easy to get matched, and the Baxandall set up gives unity gain and low Rout and high Dn reduction and is regarded as very good practice for tone controls but of course if a 12AX7 is used the effectiveness of the FB is reduced because the input series R in the FB network causes so much Dn. I have used 12AX7 in such arrangements but I have not found the phenomenon of the Dn you speak of to be a bother, but then I wasn't really looking for it, nor did I compare the Dn of a simple CC gain stage and that of one with shunt FB network to see if I got the expected reduction of open loop Dn. If one has a gain stage with open loop gain = 60 with a given RL, and then one reduces the gain to just under 1.0 with a shunt FB network which results in the same RL, then one would expect to see closed loop Dn reduced by a factor of approx 1/60, or a heck of a lot, and difficult to measure at say 0.1Vrms signal level. I have my tone control arranged so that it can be switched in or out of the signal path and nobody can tell by listening when I have it either in or out, if I have it set for a flat response. Patrick Turner. Cheers ian Cheers Ian- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#13
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Alex wrote:
Alec wrote: Out of curiosity, could you short out the plate load of 12AX7? (AC component by a 10uF from plate to ground.) If the distortion reduces, it is Miller effect amplifying the distortion of the tube. If the distortion remains about the same -- then it is some kind of weird non-linear conduction of the grid. Grid current should not appear with the bias more than --1.4...1.5V, I would think. Good idea! I'm fascinated by the idea of non-linear capacitance. A cold valve has measurable capacitance between pairs of electrodes. Considering the dielectric is a vacuum, these should be linear. When the valve is operating, gain between electrodes appears to multiply capacitance, so that if the gain is non-linear, so the capacitance also appears non-linear. So far, all this assumes that the capacitance is, itself, linear. In addition to the non-linear effects of linear capacitance, there are non-linear effects that are not frequency-dependent and that don't necessarily impinge on gain. Standing grid current is an example. Since it varies with voltage, grid current may result in distortion if it is allowed to effect grid voltage, as Ian's experiments demonstrate, maybe. So what is non-linear capacitance? Is it merely one or both of these effects, lumped together for convenience perhaps? Or is it something else? Just thinking of an obvious example of a capacitance that is said to be non-linear, such as a ceramic, where the non-linearity arises from a bulk transport of electrons in the dielectric. In a valve we have moving electrons, obviously. What is called capacitance, and what we call something else, may be no more than a point of view. And what if you reduce the heater voltage? What's your thinking? What in general is the effect of reduced heater current? My guess was that it's related to perveance, and that it should therefore make all the curves flatter. Just a guess. Also, if the grid is very close to the cathode, then reducing the cloud of thermionically-excited electrons may have a greater effect on grid current than one might otherwise expect, especially if they don't see the anode as a particularly attractive destination when Vak is low. It would be easier for me, and possibly others if others exist, if Ian would talk in terms of DC voltage and current so we don't have to translate RMS and those singularly unhelpful dB thingies. If we are to distinguish between DC and AC effects, and relate the two, we need DC and AC voltages and currents that are directly comparable. Otherwise we could eventually discover the error is in his arithmetic. Ian |
|
#14
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
"Ian Bell" wrote in message ... I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? Cheers Ian Hello Ian, Out of curiosity, could you short out the plate load of 12AX7? (AC component by a 10uF from plate to ground.) If the distortion reduces, it is Miller effect amplifying the distortion of the tube. If the distortion remains about the same -- then it is some kind of weird non-linear conduction of the grid. Grid current should not appear with the bias more than --1.4...1.5V, I would think. And what if you reduce the heater voltage? Regards, Alex |
|
#15
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 17, 6:58 am, Ian wrote: Patrick Turner wrote: On Sep 16, 5:34 am, Ian wrote: snip a bit, and now reply to the rest of your post.... It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. That's interesting. I've never tested a White follower for its distortion, but Dn should be low. I would have maybe thought a 12BH7 a little weak for headphones and a direct connection to a 32 ohm load. It is. I originally used a paralleled pair running at 30mA quiescent total CF but that soon ran out of drive current on one half cycle being single ended. That's why I decided to try the White. Are you using an OPT? Yes, it is a Sowter 8865 http://www.sowter.co.uk/specs/8665.htm The load for any kind of follower should be still at least above 2 x Ra, and BH7 has Ra = about 5k per half, so RL should be 10k minimum for 1 triode section and therefore 5k for the pair of triodes. And 5k when driven with a White follower should give very low Dn. At present I have a single 12BH7 white follower running at 12mA driving this transformer. For 1Vrms on the secondary into 34 ohms (88mW) requires less than 2.5mA rms in the primary or 7mA pp which is well within the 20mA or so current awing available. A pair of trioded EL84 make a nice white followe r with 25mA for Ia and about Ea = about 200V per tube. Load can then be about 1k, capacitor coupled off the cathode-anode joint between the two triodes. The problem right now does not seem to be in the white follower. For one of the headphones, a 100R type, it cant take up to 3V rms which implies 36V rms on the primary or nearly 102V pp. Even using a plate bootstrapped 12AX7 with a gain of around 80 I need nearly 1.3V pp or 0.65V peak at the grid - and this definitely seems to be in the grid current region for the 12AX7 even biassed at over -1.5V and as far as I can make out this is where the distortion appears to arise. Cheers Ian I did use a White follower as an output buffer in a wien bridge oscillator I once made. This buffer had a pair of EL36 or 6CM5 and its contribution to Dn was negligible. But I later thought it was simpler and just as effective to make an oscillator buffer to be the driver of the positive and negative FB networks of the wien bridge oscillator and thus get a much higher possible top frequency which is now 250kHz. The trouble with wien bridge oscillators and HF is that the R&C networks for the FB become low impedance at F goes higher and this loads an output gain pentode adversely, especially if one wants about 14Vrms of output which is so easy at 1kHz, and RC networks which hardly loads the output gain pentode. My oscillator uses a 6AC7 directly coupled to the CF output buffer. The buffer has a dc load with a CC sink, so its output mainly only sees the cap coupled FB loops and the potentiometers which adjust the output to the outside world. I've been meaning to couple together a few of the identical 3 gang radio tuning caps I have to make a much better wien bridge oscillator which will have a higher top frequency and a much lower bottom frequency. I find the stability of operation of the wien bridge oscillator is best when using radio tuning caps rather than trying to use a dual gang potentiometer. But time is my enemy. Patrick Turner. As tubes age, the grid can act strangely and its slight tendency to be always slightly more negative than the bias supply, or say 1V ess than 0V where Rg = 10M, becomes less of a tendency, and the bias control becomes lost and grid slowly becomes positive and Ia then is turned on more so that older tubes can have a positive grid with very low Ea and high Ia as a result. This happens in old radios and nobody notices until Ea has declined to about +20V only, and distortion is beginning to get audible at a percent or more. Don't think that is a problem as I am using new tubes right now. Your 12AX7 operating conditions seem about just right, and I guess changing samples of 12AX7 should not change the phenomena you see - although some well worn 'AX7 samples will probably measure more THD than newer ones with a harder vacuum. As I said both ones I tried are new - both by EH. Things could be worse - if you used a small BJT with bypassed emitter resistance - the THD will be horrendous, but with a much bigger difference between having a 100k in series with source and base and having no series R. LOL Cheers Ian Patrick Turner.- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#16
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 18, 8:39 am, Ian wrote: flipper wrote: On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: Snip for brevity. Exactly the same as at the grid - about 2H at -30dB or worse. If I short the input resistor 2H at the grid drops below -70dB and at the anode falls to -40dB which is about what I would expect. So I was expecting -40dB 2H at the anode and got a lot higher so traced back to the grid and found it was the same but that it wnet away with near zero source impedance. Your variations of Dn with input series resistance for 12AX7 is interesting, and not many ppl would be wise to it. But what of other tubes? In all my tests for line level preamps using 12AU7, 6SN7, and my favourite, 6CG7, the Dn you mention as a result of high input source resistance never seemed to occur. What if you change from 12AX7 to 12AU7? The 'AU7 should work in your circuit with minor load&biasing changes. I recall measuring a 6CG7 line stage with 100k volume control before a CC gain tube using 47k anode load then direct coupled CF to output and at typical levels to drive a power amp at normal listening levels Dn was always less than 0.02%, or about -74dB, which I thought would always be inaudible, and completely overwhelmed by the Dn of the power amp and speakers. I am tempted to use one of my 6CG7 mu followers as a driver but that would only have a gain of about 20. To get the 102V pp drive mentioned in my last post would require an input of 5.1V pp or about 2.5V peak - my 5mA 6CG7 mu followers bias at close to -5V so it should avoid grid current. I may then need another gain stage to achieve a suitable sensitivity and of course the input signal would become even greater if I use NFB to reduce the white distortion. Of course teh 6CG7 mu follower only achieve about 0.4% distortion at 20V rms output itself and I really need about twice this. I am tempted to try changing the windings of the transformer to the 6:1 tap to lower the drive requirements. Cheers Ian BTW, the amp with 6CG7 was tried with 4 different brands of 6CG7 and with music and with 4 of us listening for an afternoon. There was a considerable difference to the sound between excellent with Siemans NOS and not very good at all with Sovtek, but I thought an Aust made was right up there with the Siemans. The NOS Mullard was just polite sounding. all tubes would not have had very different Dn profiles. But I have never built a preamp with 100k of series input resistance. But then there is a another consideration. Many line stages are made using a gain tube which may or may not have a follower buffer to its outout. And there may well be a volume control at the input. and after the volume control the gain tube grid has a 100k series R to its grid and say a 560k taken to the buffer output. This is a simple shunt FB network and it gives closed loop gain of about 5x, and if the gain triode was a 12AX7 with open loop gain = 70x, then gain reduction with FB = 1/14, and any Dn is also reduced by the same margin UNLESS of course the series input resistance increases the Dn which would result in much less Dn reduction than a factor of 1/14. Baxandall tone control stages may typically have a CF input stage or low Rout gain stage driving the input to a shunt FB network around a tone control gain tube. The potentiometers used for the tone control are linear pots and easy to get matched, and the Baxandall set up gives unity gain and low Rout and high Dn reduction and is regarded as very good practice for tone controls but of course if a 12AX7 is used the effectiveness of the FB is reduced because the input series R in the FB network causes so much Dn. I have used 12AX7 in such arrangements but I have not found the phenomenon of the Dn you speak of to be a bother, but then I wasn't really looking for it, nor did I compare the Dn of a simple CC gain stage and that of one with shunt FB network to see if I got the expected reduction of open loop Dn. If one has a gain stage with open loop gain = 60 with a given RL, and then one reduces the gain to just under 1.0 with a shunt FB network which results in the same RL, then one would expect to see closed loop Dn reduced by a factor of approx 1/60, or a heck of a lot, and difficult to measure at say 0.1Vrms signal level. I have my tone control arranged so that it can be switched in or out of the signal path and nobody can tell by listening when I have it either in or out, if I have it set for a flat response. Patrick Turner. Cheers ian Cheers Ian- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#17
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 19, 1:48*am, "Ian Iveson"
wrote: Alex wrote: Alec wrote: Out of curiosity, could you short out the plate load of 12AX7? (AC component by a 10uF from plate to ground.) If the distortion reduces, it is Miller effect amplifying the distortion of the tube. If the distortion remains about the same -- then it is some kind of weird non-linear conduction of the grid. Grid current should not appear with the bias more than --1.4...1.5V, I would think. Good idea! I'm fascinated by the idea of non-linear capacitance. A cold valve has measurable capacitance between pairs of electrodes. Considering the dielectric is a vacuum, these should be linear. When the valve is operating, gain between electrodes appears to multiply capacitance, so that if the gain is non-linear, so the capacitance also appears non-linear. So far, all this assumes that the capacitance is, itself, linear. In addition to the non-linear effects of linear capacitance, there are non-linear effects that are not frequency-dependent and that don't necessarily impinge on gain. Standing grid current is an example. Since it varies with voltage, grid current may result in distortion if it is allowed to effect grid voltage, as Ian's experiments demonstrate, maybe. So what is non-linear capacitance? Is it merely one or both of these effects, lumped together for convenience perhaps? Or is it something else? Just thinking of an obvious example of a capacitance that is said to be non-linear, such as a ceramic, where the non-linearity arises from a bulk transport of electrons in the dielectric. In a valve we have moving electrons, obviously. What is called capacitance, and what we call something else, may be no more than a point of view. And what if you reduce the heater voltage? What's your thinking? What in general is the effect of reduced heater current? My guess was that it's related to perveance, and that it should therefore make all the curves flatter. Just a guess. Also, if the grid is very close to the cathode, then reducing the cloud of thermionically-excited electrons may have a greater effect on grid current than one might otherwise expect, especially if they don't see the anode as a particularly attractive destination when Vak is low. It would be easier for me, and possibly others if others exist, if Ian would talk in terms of DC voltage and current so we don't have to translate RMS and those singularly unhelpful dB thingies. If we are to distinguish between DC and AC effects, and relate the two, we need DC and AC voltages and currents that are directly comparable. Otherwise we could eventually discover the error is in his arithmetic. I think the anode-grid capacitance which appears to be a larger amount of C compared to the C when there is no gain becomes variable because of the change in gain due to changes in gm and Ra with Ia changes during each wave cycle. But the Miller effect is a shunt NFB effect and as F rises the Cag becomes a lower impedance which reduces gain. Distortion tends to be corrected by the NFB. In 12AX7 with static Ca-g = 2pF, with gain at 70 the C apparent becomes 140pF, and if you have 100k series R between source and grid then you have a LPF formed with 100k and 140pF and a pole at 11.357kHz. But then at 113kHz the Miller C is NOT 140pF but less because the gain is less. Its not as simple as it seems. Observation of a working tube and drawing the response may better answer what is going on. But in about 1932 someone put another grid between g1 and a and the C which so greatly interferred with gain as F rose was largely defeated without resorting to using additional triode tubes. If the 'AX7 anode is shunted to 0V via a 10uF cap the Cag remains at the same low static nil-signal value of maybe 2pF plus whatever other stray C and Cgk may exist. So Miller is not doing anything unless gain is present. If the input to the 12AX7 is via the cathode and with grounded grid the Miller effect is defeated but its always been a PIA to have to make a low impedeance drive to a gain tube. One way is to have a CF and gain tube set up with gain tube with grounded grid and both cathodes taken to a constant current sink, and then you have a high Z input and almost no miller effect although the Cag of the gain tube still exists between anode and the grounded grid, and so the resulting low pass filter is the summed total of the static Cag plus stray C acting as shunt C after a resistance of Ra, which is about 65kohms with 12AX7. Bandwidth of this sort of gain stage is wider than just one triode, but Dn could be greater. Then there is cascode connection in which the bottom triode has very little gain because its anode drives into the low Rin of the top tube cathode with a grounded grid. If top tube has gain 70 and RL of 140k, then Rkin = 140k / 70 = 2k, and bottom tube gain is only maybe 3, with very low Miller effect. But this gives overall gain of about 210, not bad, if gain is what you want. Cascode Dn is worse though, and if there is series R between source and G1 then the Dn effects being discussed will still appear. I recall seeing a cascode stage schematic from a magazine in 1995 using the two halves of a 12AU7 being touted as being excellent as an input stage for MM carts. I don't see why not. Patrick Turner. Ian |
|
#18
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 19, 7:14*pm, Ian Bell wrote:
Patrick Turner wrote: On Sep 17, 6:58 am, Ian *wrote: Patrick Turner wrote: On Sep 16, 5:34 am, Ian * *wrote: * snip a bit, and now reply to the rest of your post.... It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. That's interesting. I've never tested a White follower for its distortion, but Dn should be low. I would have maybe thought a 12BH7 a little weak for headphones and a direct connection to a 32 ohm load. It is. I originally used a paralleled pair running at 30mA quiescent total CF but that soon ran out of drive current on one half cycle being single ended. That's why I decided to try the White. Are you using an OPT? Yes, it is a Sowter 8865 http://www.sowter.co.uk/specs/8665.htm The load for any kind of follower should be still at least above 2 x Ra, and BH7 has Ra = about 5k per half, so RL should be 10k minimum for 1 triode section and therefore 5k for the pair of triodes. And 5k when driven with a White follower should give very low Dn. At present I have a single 12BH7 white follower running at 12mA driving this transformer. For 1Vrms on the secondary into 34 ohms (88mW) requires less than 2.5mA rms in the primary or 7mA pp which is well within the 20mA or so current awing available. A pair of trioded EL84 make a nice white followe r with 25mA for Ia and about Ea = about 200V per tube. Load can then be about 1k, capacitor coupled off the cathode-anode joint between the two triodes. The problem right now does not seem to be in the white follower. For one of the headphones, a 100R type, it cant take up to 3V rms which implies 36V rms on the primary or nearly 102V pp. Even using a plate bootstrapped 12AX7 with a gain of around 80 I need nearly 1.3V pp or 0.65V peak at the grid - and this definitely seems to be in the grid current region for the 12AX7 even biassed at over -1.5V and as far as I can make out this is where the distortion appears to arise. I'm not sure exactly what te load matches are using the Sowter OPT which seems like a like peice of kit. Sowter specs are not clear as they should be, and don't seem to indicate exactly what the primary load is with each secondary load one might choose. 102VP-P is 36Vrms, and if AX7 gain = 60, then you'd need 0.6Vrms grid input = 1.69VP-P and yes, you'd have a bother with distortion getting high. In the last Phone amp I made for a customer last year I used a single EL84 in triode with an OPT from an old Trio stereo receiver. I had 12AU7 driving the EL84, and some global NFB and a resistance divider off the 16 oms sec to drive the phones. The OPT was about 5K:16, 8 &4. Noise ended up less than could be heard with phones and the drive signal to the EL84 was below about 6Vrms at clipping which was way above what was needed for normal listening so the driver tube contributed negligible Dn. But about 12 years ago I used a 6GW8 triode pentode and tried driving the phones off the pentode section running as a CF with maybe 35mA and with the fairly high gain triode as the driver. No OPT. But its Dn was much higher. Not bad though; the guy wanted this as part of a head amp for guitar sound monitoring without having to use a speaker. There must be many ways to do a headphone amp. An SE mosfet with CCS dc load makes a good driver for phones with no OPT. Patrick Turner |
|
#19
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 19, 7:22*pm, Ian Bell wrote:
Patrick Turner wrote: On Sep 18, 8:39 am, Ian *wrote: flipper wrote: On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: Snip for brevity. Exactly the same as at the grid - about 2H at -30dB or worse. If I short the input resistor 2H at the grid drops below -70dB and at the anode falls to -40dB which is about what I would expect. So I was expecting -40dB 2H at the anode and got a lot higher so traced back to the grid and found it was the same but that it wnet away with near zero source impedance. Your variations of Dn with input series resistance for 12AX7 is interesting, and not many ppl would be wise to it. But what of other tubes? In all my tests for line level preamps using 12AU7, 6SN7, and my favourite, 6CG7, the Dn you mention as a result of high input source resistance never seemed to occur. What if you change from 12AX7 to 12AU7? The 'AU7 should work in your circuit with minor load&biasing changes. I recall measuring a 6CG7 line stage with 100k volume control before a CC gain tube using 47k anode load then direct coupled CF to output and at typical levels to drive a power amp at normal listening levels Dn was always less than 0.02%, or about -74dB, which I thought would always be inaudible, and completely overwhelmed by the Dn of the power amp and speakers. I am tempted to use one of my 6CG7 mu followers as a driver but that would only have a gain of about 20. To get the 102V pp drive mentioned in my last post would require an input of 5.1V pp or about 2.5V peak - my 5mA 6CG7 mu followers bias at close to -5V so it should avoid grid current. I may then need another gain stage to achieve a suitable sensitivity and of course the input signal would become even greater if I use NFB to reduce the white distortion. Of course teh 6CG7 mu follower only achieve about 0.4% distortion at 20V rms output itself and I really need about twice this. I am tempted to try changing the windings of the transformer to the 6:1 tap to lower the drive requirements. Cheers Ian The use of a cascade circuit with 12AU7 and normal high ratio OPT with trioded EL84 seemed to work for me to give enough gain and allow some global NFB. I gave two lots of L&R outputs with different levels and with a DACT source select switch and attenuator. He says it makes a mighty fine preamp for his power amps. The EL84 with 5K load has a damping factor of 2 without NFB. Gain between EL84 and sec is about unity, like a follower, but the load match is much better than having say 600 ohms driven by a follower. I didn't seem to ever need to have high signal voltages anywhere. Certainly never 102VP-P. The guy runs Sennheiser phones which I found rather good, and I just used some El-cheapo crap phones which were less sensitive for the testing phase. Patrick Turner. |
|
#20
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 19, 7:14 pm, Ian wrote: Patrick Turner wrote: On Sep 17, 6:58 am, Ian wrote: Patrick Turner wrote: On Sep 16, 5:34 am, Ian wrote: snip a bit, and now reply to the rest of your post.... It is a phones amp - bootstrapped 12AX7 CC feeding a pair of 12BH7 arranged as White follower. rIght now the White produces less distortion than the 12AX7. That's interesting. I've never tested a White follower for its distortion, but Dn should be low. I would have maybe thought a 12BH7 a little weak for headphones and a direct connection to a 32 ohm load. It is. I originally used a paralleled pair running at 30mA quiescent total CF but that soon ran out of drive current on one half cycle being single ended. That's why I decided to try the White. Are you using an OPT? Yes, it is a Sowter 8865 http://www.sowter.co.uk/specs/8665.htm The load for any kind of follower should be still at least above 2 x Ra, and BH7 has Ra = about 5k per half, so RL should be 10k minimum for 1 triode section and therefore 5k for the pair of triodes. And 5k when driven with a White follower should give very low Dn. At present I have a single 12BH7 white follower running at 12mA driving this transformer. For 1Vrms on the secondary into 34 ohms (88mW) requires less than 2.5mA rms in the primary or 7mA pp which is well within the 20mA or so current awing available. A pair of trioded EL84 make a nice white followe r with 25mA for Ia and about Ea = about 200V per tube. Load can then be about 1k, capacitor coupled off the cathode-anode joint between the two triodes. The problem right now does not seem to be in the white follower. For one of the headphones, a 100R type, it cant take up to 3V rms which implies 36V rms on the primary or nearly 102V pp. Even using a plate bootstrapped 12AX7 with a gain of around 80 I need nearly 1.3V pp or 0.65V peak at the grid - and this definitely seems to be in the grid current region for the 12AX7 even biassed at over -1.5V and as far as I can make out this is where the distortion appears to arise. I'm not sure exactly what te load matches are using the Sowter OPT which seems like a like peice of kit. Sowter specs are not clear as they should be, and don't seem to indicate exactly what the primary load is with each secondary load one might choose. You are right - Sowter is generally lax in giving details of its specs - that said they do make first class transformers and this one has a primary inductance of 137H which should be more than adequate. 102VP-P is 36Vrms, and if AX7 gain = 60, then you'd need 0.6Vrms grid input = 1.69VP-P and yes, you'd have a bother with distortion getting high. In the last Phone amp I made for a customer last year I used a single EL84 in triode with an OPT from an old Trio stereo receiver. I had 12AU7 driving the EL84, and some global NFB and a resistance divider off the 16 oms sec to drive the phones. The OPT was about 5K:16, 8&4. Noise ended up less than could be heard with phones and the drive signal to the EL84 was below about 6Vrms at clipping which was way above what was needed for normal listening so the driver tube contributed negligible Dn. It is good to know that an SE triode EL84 works OK. Sowter also make an SE phones transformer but it is a huge expensive monster (type 9351) http://www.sowter.co.uk/headphone-transformer.php I think right now that is my last resort but EL84 based white follower seems an interesting option. Cheers Ian But about 12 years ago I used a 6GW8 triode pentode and tried driving the phones off the pentode section running as a CF with maybe 35mA and with the fairly high gain triode as the driver. No OPT. But its Dn was much higher. Not bad though; the guy wanted this as part of a head amp for guitar sound monitoring without having to use a speaker. There must be many ways to do a headphone amp. An SE mosfet with CCS dc load makes a good driver for phones with no OPT. Patrick Turner |
|
#21
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 19, 7:22 pm, Ian wrote: Patrick Turner wrote: On Sep 18, 8:39 am, Ian wrote: flipper wrote: On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: Snip for brevity. Exactly the same as at the grid - about 2H at -30dB or worse. If I short the input resistor 2H at the grid drops below -70dB and at the anode falls to -40dB which is about what I would expect. So I was expecting -40dB 2H at the anode and got a lot higher so traced back to the grid and found it was the same but that it wnet away with near zero source impedance. Your variations of Dn with input series resistance for 12AX7 is interesting, and not many ppl would be wise to it. But what of other tubes? In all my tests for line level preamps using 12AU7, 6SN7, and my favourite, 6CG7, the Dn you mention as a result of high input source resistance never seemed to occur. What if you change from 12AX7 to 12AU7? The 'AU7 should work in your circuit with minor load&biasing changes. I recall measuring a 6CG7 line stage with 100k volume control before a CC gain tube using 47k anode load then direct coupled CF to output and at typical levels to drive a power amp at normal listening levels Dn was always less than 0.02%, or about -74dB, which I thought would always be inaudible, and completely overwhelmed by the Dn of the power amp and speakers. I am tempted to use one of my 6CG7 mu followers as a driver but that would only have a gain of about 20. To get the 102V pp drive mentioned in my last post would require an input of 5.1V pp or about 2.5V peak - my 5mA 6CG7 mu followers bias at close to -5V so it should avoid grid current. I may then need another gain stage to achieve a suitable sensitivity and of course the input signal would become even greater if I use NFB to reduce the white distortion. Of course teh 6CG7 mu follower only achieve about 0.4% distortion at 20V rms output itself and I really need about twice this. I am tempted to try changing the windings of the transformer to the 6:1 tap to lower the drive requirements. Cheers Ian The use of a cascade circuit with 12AU7 and normal high ratio OPT with trioded EL84 seemed to work for me to give enough gain and allow some global NFB. Cascade should give enough gain but adds to the zeros in the loop so NFB stability is compromised - not a route I want to take unless I absolutely have to. I gave two lots of L&R outputs with different levels and with a DACT source select switch and attenuator. He says it makes a mighty fine preamp for his power amps. The EL84 with 5K load has a damping factor of 2 without NFB. Gain between EL84 and sec is about unity, like a follower, but the load match is much better than having say 600 ohms driven by a follower. I didn't seem to ever need to have high signal voltages anywhere. Certainly never 102VP-P. The guy runs Sennheiser phones which I found rather good, and I just used some El-cheapo crap phones which were less sensitive for the testing phase. Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 541V pp is less of a problem than the 3600R load. His other phones are 34R types. For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. So I think that where I need to head. The problem still remains how to get the 60V pp swing without unnecessary first stage distortion. Cheers Ian Patrick Turner. |
|
#22
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
"Ian Iveson" wrote in message ... Alex wrote: Alec wrote: Out of curiosity, could you short out the plate load of 12AX7? (AC component by a 10uF from plate to ground.) If the distortion reduces, it is Miller effect amplifying the distortion of the tube. If the distortion remains about the same -- then it is some kind of weird non-linear conduction of the grid. Grid current should not appear with the bias more than --1.4...1.5V, I would think. Good idea! I'm fascinated by the idea of non-linear capacitance. A cold valve has measurable capacitance between pairs of electrodes. Considering the dielectric is a vacuum, these should be linear. When the valve is operating, gain between electrodes appears to multiply capacitance, so that if the gain is non-linear, so the capacitance also appears non-linear. So far, all this assumes that the capacitance is, itself, linear. In addition to the non-linear effects of linear capacitance, there are non-linear effects that are not frequency-dependent and that don't necessarily impinge on gain. Standing grid current is an example. Since it varies with voltage, grid current may result in distortion if it is allowed to effect grid voltage, as Ian's experiments demonstrate, maybe. So what is non-linear capacitance? Is it merely one or both of these effects, lumped together for convenience perhaps? Or is it something else? Just thinking of an obvious example of a capacitance that is said to be non-linear, such as a ceramic, where the non-linearity arises from a bulk transport of electrons in the dielectric. In a valve we have moving electrons, obviously. What is called capacitance, and what we call something else, may be no more than a point of view. And what if you reduce the heater voltage? What's your thinking? What in general is the effect of reduced heater current? My guess was that it's related to perveance, and that it should therefore make all the curves flatter. Just a guess. Also, if the grid is very close to the cathode, then reducing the cloud of thermionically-excited electrons may have a greater effect on grid current than one might otherwise expect, especially if they don't see the anode as a particularly attractive destination when Vak is low. It would be easier for me, and possibly others if others exist, if Ian would talk in terms of DC voltage and current so we don't have to translate RMS and those singularly unhelpful dB thingies. If we are to distinguish between DC and AC effects, and relate the two, we need DC and AC voltages and currents that are directly comparable. Otherwise we could eventually discover the error is in his arithmetic. Ian Grid-to-plate capacitance is pretty linear. However, since the gain is non-linear, I thought that the Miller effect related virtual capacitance referred to grid also becomes non-linear. But there is another idea which crossed my mind. Cathode-to-grid capacitance is inherently non-linear. It is a known fact. In fact, a vacuum diode (or cathode-to-grid section of a triode) works as a varicap or varactor. If the negative bias is high, then the electon cloud is all kept at bay at the cathode surface. When bias voltage reduces, hotter electrons are able to travel closer to the grid (or diode plate). The electron cloud is sort of a conductive media. Thus the cloud comes closer to the grid wires, increasing grid-to-cathode capacitance. This varactor effect at the grid is always there regardless of the plate load. Looking forward to Ian Bell's test results. |
|
#23
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 20, 5:46*am, Ian Bell wrote:
Patrick Turner wrote: On Sep 19, 7:14 pm, Ian *wrote: Patrick Turner wrote: On Sep 17, 6:58 am, Ian * *wrote: Patrick Turner wrote: On Sep 16, 5:34 am, Ian * * *wrote: * *snip a bit, You are right - Sowter is generally lax in giving details of its specs - that said they do make first class transformers and this one has a primary inductance of 137H which should be more than adequate. 137H is entirely adequate for primary L. It isn't too hard if the core µ is high, which I suspect it is because it is mu-metal. It saturates easily with DC but with only AC it is very good for low signal levels. Last year I had a long session of emails with Brian Sowter about him making replacement OPT for Quad-II amps. I gave him my design details and wanted a quote. But he didn't want to wind it how I specified and I asked about exactly how he would do it and he declined to give any details, and he said he'd do it his way only. So I gently insisted what his general principles would be regarding the bobbin layers and where he'd put the 10% cathode feedback windings and he suggested he'd bifilar wind them at the anode ends of the anode windings. I replied I would NEVER do any such thing, and would want the transformers wound as I specified or not at all, and that his method resulted in likely arcing between the anode windings and cathode FB windings and that the bifilar arrangement where said it ought to go would cause a huge unwanted increase in shunt capacitance between the anode and 0V. He did not reply further. The emails were quite friendly of course but occurred during a world cruise he was doing, so there's money in transformers. But I suspect he may be getting on bit maybe towards retirement. But so am I, and yet I relish ideas different to my own if they seem to be good ones. If you Google Sowter history, you'll find an older Mr Sowter who was a leading expert on transformers way back in the 1930s. Perhaps this famous Sowter was Brian's dad; I don't know, but the existing company probably has a huge pile of their designs for 1,001 types of transfromers and when some Colonial Upstart from Australia comes along to tell a big company how it ought to wind its OPTs then they don't warm to the advice. Anyway, mehinks Brian doesn't know as much as his ancestors. I cannot ever proceed to hire anyone on the basis of "trust us we've been operating for 70 years". I need to know ALL the grisly details. 102VP-P is 36Vrms, and if AX7 gain = 60, then you'd need 0.6Vrms grid input = 1.69VP-P and yes, you'd have a bother with distortion getting high. In the last Phone amp I made for a customer last year I used a single EL84 in triode with an OPT from an old Trio stereo receiver. I had 12AU7 driving the EL84, and some global NFB and a resistance divider off the 16 oms sec to drive the phones. The OPT was about 5K:16, 8&4. Noise ended up less than could be heard with phones and the drive signal to the EL84 was below about 6Vrms at clipping which was way above what was needed for normal listening so the driver tube contributed negligible Dn. It is good to know that an SE triode EL84 works OK. Sowter also make an SE phones transformer but it is a huge expensive monster (type 9351) It does not need to be huge. The OPT I used were very generic 5W types meant for EL84 in pentode mode, and have plain GOSS with air gap, and they saturate at 50Hz at full power which is bloody awful, but then Trio was just another cheap **** brand and transformer weight size and cost was never a priority. The core was a 20mm stack of 20mm wasteless material with Lp maybe less than 20H. But they function well with the low levels which are well away from saturation when using headphones. EL86 is another pentode which makes a good little triode with Ra = 1.4k and gain about 9 and it is best with Ea = 200V approx and maybe Ia = 40mA. But you could use EL86 with a 300V supply with Ea = 150V approx to make a White follower and with Ia = 40mA and get +/- 0.64VP- P into 32 ohms without anything clipping. Its not going to be as linear as a load of 320ohms or 1k which would be much better. If you don't know what the primary load is it is easily found with a series 10R between cathode and OPT primary when loaded by the secondary load. Primary load = Cathode output signal voltage / current measured in 10R. http://www.sowter.co.uk/headphone-transformer.php I think right now that is my last resort but EL84 based white follower seems an interesting option. I recall that the White CF does have advantages - it allows PP operation and thus 2H cancelling, and output load regulation via the current flow in the R above the top triode. If the load is low, the load current is high and the signal produced in the top anode R is then transfered to the botom triode to make it work harder. Its not an ideal way of doing things but is OK over a range of loads. The Rout of a White follower using a twin triode is slightly lower than if you simply paralleled the two triodes ina plain CF which of course can be made to work optimally using a CCS for the dc load sink. The plain CF has only marginally higher Rout. Patrick Turner. |
|
#24
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 20, 6:01*am, Ian Bell wrote:
Patrick Turner wrote: On Sep 19, 7:22 pm, Ian *wrote: Patrick Turner wrote: On Sep 18, 8:39 am, Ian * *wrote: flipper wrote: On Fri, 17 Sep 2010 10:48:50 +0100, Ian wrote: flipper wrote: On Thu, 16 Sep 2010 21:58:37 +0100, Ian wrote: Snip for brevity. The use of a cascade circuit with 12AU7 and normal high ratio OPT with trioded EL84 seemed to work for me to give enough gain and allow some global NFB. Cascade should give enough gain but adds to the zeros in the loop so NFB stability is compromised - not a route I want to take unless I absolutely have to. You may find you only need about 15dB of global NFB and this is easy enough to stabilise because you have only two stages. The White follower is a case of a lot of local series voltage FB with current regulation added. It can be unstable. I gave two lots of L&R outputs with different levels and with a DACT source select switch and attenuator. He says it makes a *mighty fine preamp for his power amps. The EL84 with 5K load has a damping factor of 2 without NFB. Gain between EL84 and sec is about unity, like a follower, but the load match is much better than having say 600 ohms driven by a follower. I didn't seem to ever need to have high signal voltages anywhere. Certainly never 102VP-P. The guy runs Sennheiser phones which I found rather good, and I just used some El-cheapo crap phones which were less sensitive for the testing phase. Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 51V pp is less of a problem than the 3600R load. OK, I wasn't sure what loads you actually have. 12:1 turn ratio is 144:1 Z ratio which is 14.4k :100 ohms, or 4.8k : 33 ohms. You could use the 12BH7 as a parallled unit to make a triode with Ra = 2k5 approx and µ = 17. For 36Vrms into 14k the BH7 gain = 14, and you'd need only 2.6Vrms to the BH7 grids. The anode Vswing is above and below the Ea voltage and could easily be Much higher than 36Vrms before any clipping. His other phones are 34R types. *For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. So I think that where I need to head. The problem still remains how to get the 60V pp swing without unnecessary first stage distortion. The EL84 or 6V6 was a common tube used in any situation where headphones were to be used at the output from say a communications receiver. Usually you could have a small speaker for a couple of watts max. The generous power capability meant that maybe you could have 4Vrms into 8 ohms, and at say 3% THD, but at 0.4Vrms the power into 32 ohms was only 5mW but plenty and THD would be 0.01%. Noise is usually the problem to overcome with good hi-fi phone amps. So even though the load could be a speaker of 8 ohms off the sec of the OPT, you'd never want to connect any phones to the speaker terminals because the noise usually is way too high and so you'd need to have at least a 4:1 resistance divider, say 22 ohms plus 4.7 ohms and drive the phones with the signal across the 4.7 ohms regardless of the phones impedance. I've done this in a mod to a GlowOne amp made in china and with EL84 outputs but which gave much too much noise without the resistance divider. Ming-Da phone amps also suffer the same lack of design finesse. The resistance divider allows the primary signal to be high and into a high primary load because the secondary is say about 24 ohms even though the tube is happy with 8 ohms. The P load is then quite a high load where THD becomes negligible and the SNR with R divider is improved nicely about 12dB, enough to get the tube across the line as a valid low noise phone amp. So where one had an OPT to work off the anode circuit of a triode meant to drive 5k as the lowest load and be able to drive 8 ohms then the TR = 25:1, and for headphone drive without a resistance divider the use of a higher ratio OPT would seem best practice so taps giving 50:1 or 100:1 would seem appropriate, if the response bandwidth can be maintained. Last year tried to convince the US based maker of the GlowOne amps who had asked me for his advice after a customer here in Canberra complained about the shortcomings of his product. There were several mods needed and I don't know whether they are now a feature of production but the US based maker was a young chirpy lawyer who was second generation chinese who had connections "back home" so he could dabble in making some little product and make an extra buck. But he didn't understand my advice very much. There are factories in China who will make anything you like with tubes and do the design work then print any old groovy name on the front and oh how the dough rolls in. Its mainly rather junky stuff that is made this way, ie, poor copies of circuits from the 1950s but presented with good looks. But I digress... The White follower is rather a good thing where the signal levels are low and loads high and where you want a "scientifically pure outcome" with very little THD as much as it is possible when using tubes and in a somewhat simple arrangement. Patrick Turner. Cheers Ian Patrick Turner.- Hide quoted text - - Show quoted text -- Hide quoted text - - Show quoted text - |
|
#25
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 541V pp is less of a problem than the 3600R load. His other phones are 34R types. For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. Doesn't this argument omit the issue of efficiency, part of the matter of sensitivity that you originally acknowledged? If the higher impedance 'phones are proportionately more efficient, then you might be home and dry. Or not. Alternatively, it's reasonable to consider the old-school 100R to be for gentle music, and the 34R for rock. Tell him this is why god gave him two 'phones. My bet is he'll be more than satisfied with what you've already done. What, incidentally, if he blows his ears out? Aren't there some active safety guidlines for manufacturers? Could he sue? Ian |
|
#26
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 20, 5:46 am, Ian wrote: Patrick Turner wrote: On Sep 19, 7:14 pm, Ian wrote: Patrick Turner wrote: On Sep 17, 6:58 am, Ian wrote: Patrick Turner wrote: On Sep 16, 5:34 am, Ian wrote: snip a bit, You are right - Sowter is generally lax in giving details of its specs - that said they do make first class transformers and this one has a primary inductance of 137H which should be more than adequate. 137H is entirely adequate for primary L. It isn't too hard if the core µ is high, which I suspect it is because it is mu-metal. It saturates easily with DC but with only AC it is very good for low signal levels. Last year I had a long session of emails with Brian Sowter about him making replacement OPT for Quad-II amps. I gave him my design details and wanted a quote. But he didn't want to wind it how I specified and I asked about exactly how he would do it and he declined to give any details, and he said he'd do it his way only. So I gently insisted what his general principles would be regarding the bobbin layers and where he'd put the 10% cathode feedback windings and he suggested he'd bifilar wind them at the anode ends of the anode windings. I replied I would NEVER do any such thing, and would want the transformers wound as I specified or not at all, and that his method resulted in likely arcing between the anode windings and cathode FB windings and that the bifilar arrangement where said it ought to go would cause a huge unwanted increase in shunt capacitance between the anode and 0V. He did not reply further. The emails were quite friendly of course but occurred during a world cruise he was doing, so there's money in transformers. But I suspect he may be getting on bit maybe towards retirement. But so am I, and yet I relish ideas different to my own if they seem to be good ones. If you Google Sowter history, you'll find an older Mr Sowter who was a leading expert on transformers way back in the 1930s. Perhaps this famous Sowter was Brian's dad; I don't know, but the existing company probably has a huge pile of their designs for 1,001 types of transfromers and when some Colonial Upstart from Australia comes along to tell a big company how it ought to wind its OPTs then they don't warm to the advice. Anyway, mehinks Brian doesn't know as much as his ancestors. I cannot ever proceed to hire anyone on the basis of "trust us we've been operating for 70 years". I need to know ALL the grisly details. Interesting story ( I was aware of the original MR. Sowter and I think I have an AES paper of his about audio transformer design. As you say, Brian is always very friendly. He has always answered my questions no matter how technical but I admit to not having the skills to TELL him how I think he should wind a transformer. 102VP-P is 36Vrms, and if AX7 gain = 60, then you'd need 0.6Vrms grid input = 1.69VP-P and yes, you'd have a bother with distortion getting high. In the last Phone amp I made for a customer last year I used a single EL84 in triode with an OPT from an old Trio stereo receiver. I had 12AU7 driving the EL84, and some global NFB and a resistance divider off the 16 oms sec to drive the phones. The OPT was about 5K:16, 8&4. Noise ended up less than could be heard with phones and the drive signal to the EL84 was below about 6Vrms at clipping which was way above what was needed for normal listening so the driver tube contributed negligible Dn. It is good to know that an SE triode EL84 works OK. Sowter also make an SE phones transformer but it is a huge expensive monster (type 9351) It does not need to be huge. The OPT I used were very generic 5W types meant for EL84 in pentode mode, and have plain GOSS with air gap, and they saturate at 50Hz at full power which is bloody awful, but then Trio was just another cheap **** brand and transformer weight size and cost was never a priority. The core was a 20mm stack of 20mm wasteless material with Lp maybe less than 20H. But they function well with the low levels which are well away from saturation when using headphones. That is interesting. Does anyone still make little OP transformers like this? EL86 is another pentode which makes a good little triode with Ra = 1.4k and gain about 9 and it is best with Ea = 200V approx and maybe Ia = 40mA. But you could use EL86 with a 300V supply with Ea = 150V approx to make a White follower and with Ia = 40mA and get +/- 0.64VP- P into 32 ohms without anything clipping. Its not going to be as linear as a load of 320ohms or 1k which would be much better. If you don't know what the primary load is it is easily found with a series 10R between cathode and OPT primary when loaded by the secondary load. Primary load = Cathode output signal voltage / current measured in 10R. http://www.sowter.co.uk/headphone-transformer.php I think right now that is my last resort but EL84 based white follower seems an interesting option. I recall that the White CF does have advantages - it allows PP operation and thus 2H cancelling, and output load regulation via the current flow in the R above the top triode. If the load is low, the load current is high and the signal produced in the top anode R is then transfered to the botom triode to make it work harder. Its not an ideal way of doing things but is OK over a range of loads. The Rout of a White follower using a twin triode is slightly lower than if you simply paralleled the two triodes ina plain CF which of course can be made to work optimally using a CCS for the dc load sink. The plain CF has only marginally higher Rout. Agreed. As I saif before, I don't think my problem is in the output stage but more in the driver. I am still experimenting with the 12AX7 but the results are still confusing. I think I might just build a little rig with a 12AX7 on its own to get a feel for what is actually going on.# In the meantime I am considering replacing it with a pentode - perhaps an EF86 - that should give me plenty of open loop gain and op swing - what do you think? Cheers Ian Patrick Turner. |
|
#27
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Iveson wrote:
Ian Bell wrote: Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 541V pp is less of a problem than the 3600R load. His other phones are 34R types. For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. Doesn't this argument omit the issue of efficiency, part of the matter of sensitivity that you originally acknowledged? If the higher impedance 'phones are proportionately more efficient, then you might be home and dry. Or not. Depends. I define sensitivity as dB(SPL) per mW of power to the headphone - in other words, how much noise power you get for unit electrical power. In that case, efficiency is included. It just so happens that both types of headphones produce pretty much the same sound pressure per mW electrical input and if anything the 34 ohm ones are slightly more sensitive. Alternatively, it's reasonable to consider the old-school 100R to be for gentle music, and the 34R for rock. Tell him this is why god gave him two 'phones. My bet is he'll be more than satisfied with what you've already done. Not an option at present - he wants to listen to music of any kind on either phones. What, incidentally, if he blows his ears out? Aren't there some active safety guidelines for manufacturers? Could he sue? That had occurred to me but I suspect it is the old health and safety thing where it is sufficient to say don't turn the wick up too high - and surely a similar onus must be on the phones manufacturer? Cheers ian Ian |
|
#28
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 541V pp is less of a problem than the 3600R load. His other phones are 34R types. For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. Doesn't this argument omit the issue of efficiency, part of the matter of sensitivity that you originally acknowledged? If the higher impedance 'phones are proportionately more efficient, then you might be home and dry. Or not. Depends. I define sensitivity as dB(SPL) per mW of power to the headphone - in other words, how much noise power you get for unit electrical power. In that case, efficiency is included. It just so happens that both types of headphones produce pretty much the same sound pressure per mW electrical input and if anything the 34 ohm ones are slightly more sensitive. OK, it's just that you didn't say they were equally sensitive, or efficient. In which case, OPTs with the correct ratios and optimised for the task should make both phones look equal to the amp, and the amp should look appropriately equal to both phones, no? Damping factor, power, voltage, current, etc. all come out in the wash don't they? Aren't Sowter's trannies the same price with windings of your own spec? They were with mine. Alternatively, it's reasonable to consider the old-school 100R to be for gentle music, and the 34R for rock. Tell him this is why god gave him two 'phones. My bet is he'll be more than satisfied with what you've already done. Not an option at present - he wants to listen to music of any kind on either phones. Tell him he can't have everything, and your compromise is better than the alternatives. What, incidentally, if he blows his ears out? Aren't there some active safety guidelines for manufacturers? Could he sue? That had occurred to me but I suspect it is the old health and safety thing where it is sufficient to say don't turn the wick up too high - and surely a similar onus must be on the phones manufacturer? Possibly, but there are EC guidelines on headphone outputs. Even if they are not statutory requirements, flouting them could count against you because it can be construed as reckless. I considered selling a head amp but meeting EC and CE requirements seemed daunting. I thought of limiting, simply by using zeners on the output or by using an expander in the gfb, but that can't be done practically without SS tarnishing the allure of classic design. Ian |
|
#29
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Iveson wrote:
Ian Bell wrote: Indeed, one of the really big questions is just how load does the client want the phones to be - that and the sensitivity of the phones makes a big difference to the output requirements. As a rule, 100mW into almost any headphone will blow your ears off and at least 10mW is needed to give a decent level even in the most sensitive phones. That's a 10dB difference in power that translates in to a voltage range of about 3 to 1. For this client the worst case is probably his 100R phones that will take 3V rms. That means with the 12:1 tap the primary must supply 102V pp into 14K or so. Alternatively with the 6:1 tap we need 51V pp into 3600R for about 90mW in the phones. The 541V pp is less of a problem than the 3600R load. His other phones are 34R types. For 90mW these need 1.75V on the secondary which at 12:1 is about 60V pp- again not too bad, this time into 4896R. Doesn't this argument omit the issue of efficiency, part of the matter of sensitivity that you originally acknowledged? If the higher impedance 'phones are proportionately more efficient, then you might be home and dry. Or not. Depends. I define sensitivity as dB(SPL) per mW of power to the headphone - in other words, how much noise power you get for unit electrical power. In that case, efficiency is included. It just so happens that both types of headphones produce pretty much the same sound pressure per mW electrical input and if anything the 34 ohm ones are slightly more sensitive. OK, it's just that you didn't say they were equally sensitive, or efficient. In which case, OPTs with the correct ratios and optimised for the task should make both phones look equal to the amp, and the amp should look appropriately equal to both phones, no? Damping factor, power, voltage, current, etc. all come out in the wash don't they? Aren't Sowter's trannies the same price with windings of your own spec? They were with mine. Alternatively, it's reasonable to consider the old-school 100R to be for gentle music, and the 34R for rock. Tell him this is why god gave him two 'phones. My bet is he'll be more than satisfied with what you've already done. Not an option at present - he wants to listen to music of any kind on either phones. Tell him he can't have everything, and your compromise is better than the alternatives. What, incidentally, if he blows his ears out? Aren't there some active safety guidelines for manufacturers? Could he sue? That had occurred to me but I suspect it is the old health and safety thing where it is sufficient to say don't turn the wick up too high - and surely a similar onus must be on the phones manufacturer? Possibly, but there are EC guidelines on headphone outputs. Even if they are not statutory requirements, flouting them could count against you because it can be construed as reckless. I considered selling a head amp but meeting EC and CE requirements seemed daunting. I thought of limiting, simply by using zeners on the output or by using an expander in the gfb, but that can't be done practically without SS tarnishing the allure of classic design. (fortunately) this client is in the US so CE does not apply but I think you are right - |I'll ook into it. Cheers ian Ian |
|
#30
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
I am having a strange problem with what appears to be distortion due to grid current in a 12AX7. It's a CC stage with a 3K Rk bypassed about 120K in the plate and a 300V supply. Anode sits at just over 200V and cathode just below 2V. I would not expect to see any distortion due to grid current with an input signal of 250mV rms at 2KHz but I do. There's a 1Meg from grid to ground and the test oscillator is fed to the grid via a 100K resistor. Analyzer is connected across grid to gnd. Tube off, there is no distortion with the 100K in cct or shorted. Tube powered up I get about -33dB 2H measured at the grid, -39dB 3H, -59dB 4H. Short the 100K and the 2H drops to to below -70dB. I tried changing the tube but the result is the same. What is going on? Cheers Ian I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. Cheers Ian |
|
#31
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
"Ian Bell" wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 44x d 3.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent. http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio |
|
#32
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current, typo's
"GRe" wrote in message ... "Ian Bell" wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 36x typo, should be 72x d 3.6% typo, should be 2.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent. http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio |
|
#33
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sep 24, 6:34*pm, "GRe" wrote:
"Ian Bell" wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra * 220K R'g1 680K (load) Rk * 2K2 Ia * 0.63mA Vout 36Veff A * *44x d * *3.6% Rg * 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent.http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio From the above list of performance figures we can see that there is about 1% THD per 10Vrms of output. But what is the series input resistance and to what extent would series source resistance of 100k contribute to the distortion figure of 3.6% at 36vrms output? Probably the first 10Vrms of output has much less Dn and it more rapidly increases as the grid current is a threshold effect and tends to prevent gid input voltage from going +ve so that the tube then generates 3H because of flattening of +ve and -ve wave peaks from cut off effects and input limiting combined. If the source resistance cannot be avoided at the input and if that increases THD too much, then a cathode follower input right in front of the gain tube input should lower source resistance and distortion and lessen Miller effect and increase bandwidth. A cascode input should also do the same thing using a high gm j-fet like 2Sk369, but it is considered as another whole stage, and for the fet to work you need Ia at about 5mA, so one should use a 6CG7/12AU7 paralleled. The gain of the cascode with source fully bypassed is always the product of j-fet gm x RL of the tube, and with Ia = 5mA, RL = 27k in parallel with following Rg so say about 20k. The 2sK369 has Gm = 40mA/V at 5mA, so gain = 0.04 x 20,000 = 800, which is very excessive. But this can be tamed by using a suitably sized source resistance to reduce gain to about the same as the 12AX7, ie, about say 60, and this gives a reduction factor of 60/800 = 0.075 so that if Dn was say 4% without any current FB then with CFB it becomes 0.3% at the same output voltage. The low RL value of 20k determines the Rout of the gain stage driving the White output stage. The Miller effect is negated when you have a cascode stage but there is capacitance betwen gate and source. Gate to drain C is low because gain of fet itself is low due to current FB. If the overall of cascode gain = 60, and say a 6CG7 has gain of 15 sitting on top of a fet, then the fet only needs an overall gain of 4 to give a combined total of 60. For 36Vo, one needs about 2.4Vrms at the cathode, and 0.6Vrms at the gate with about 0.033Vrms between gate and source and source resistance of around 560R taken to a well filtered negative rail with gate biased at 0V like a tube. The 560R will give good Vdc regulation. The j-fet will give much lower noise levels than any tube which is a benefit with any headphone amp. Global NFB could also be used because the cascode is a direct coupled circuit with no extra RC coupling so stability should be easy to achieve. In Allen Wright's 1988 FVP he used a cascode phono input stage with global NFB for RIAA eq, and the following line level preamp also was a j-fet + tube cascode amp which I found had far too much gain which needed revision. The cascode stage has interesting possibilities. Patrick Turner. |
|
#34
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current, typo's
GRe wrote:
wrote in message ... "Ian wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 36x typo, should be 72x d 3.6% typo, should be 2.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent. http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio Yes, that is identical to the Mullard data sheet I have.. Cheers ian |
|
#35
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Patrick Turner wrote:
On Sep 24, 6:34 pm, wrote: "Ian wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 44x d 3.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent.http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio From the above list of performance figures we can see that there is about 1% THD per 10Vrms of output. But what is the series input resistance and to what extent would series source resistance of 100k contribute to the distortion figure of 3.6% at 36vrms output? And the answer is as follows: I used 200K in the anode and 2K2 in the cathode. HT = 313V, Plate = 170V, cathode = 1.407V With 125mV rms on the grid (2KHz) via 100K I get: 2H -59dB, 3H -77dB 4H unmeasurable The signal voltage at the plate was 10V rms which means the stage gain was 80. At the plate the measured distortion was: 2H -49dB, 3H -75dB, 4H unmeasurable So the tube itself mainly contributes 10dB of 2H The input signal was then changed to give 30V rms at the plate. Plate signal distortion was then: 2H -34dB, 3H -43dB, 4H -49dB, 5H -50dB, 6H -55dB, 7H -58dB And under the same conditions looking at the transformer secondary into 100R we get: 2H -34dB, 3H -41dB, 4H -48dB, 5H -49dB, 6H -53dB, 7H -56dB Which means the white follower plus the transformer is only contributing a couple of dB to the distortion. Also, at this point the signal voltage on the grid was just under 380mV rms and the distortion at the grid was: 2H -49, 3H -62, 4H -70, 5H -77, 6H unmeasurable which shows we are approaching grid distortion but the tube distortion is still dominant. As I mentioned before, the reason for the 100K series input resistor so I can apply shunt/shunt NFB. Connecting 2Meg from the white output back the the 12AX7 grid drops the output from 2.5V rms to 0.6V rms i.e. there's about 12dB of NFB. I increased the input to give 2.5V rms at the transformer secondary and the distortion at the transformer secondary was then: 2H-45dB, 3H -50dB, 4H -55dB, 6H -58dB, 7H -61dB which is about 11dB lower overall as expected. So, this basically seems to work. At 30V rms the 12AX7 produces about 2% distortion (-34dB) which compares well with the data sheet figure of 2.5% at 367V rms. The White follower and transformer do not seem to add significantly to this. NFB gives the expected overall reduction to just under 0.6%. I would still like the distortion to be lower so I may well try the EF86. I am not sure how much there is to be gained. From the data sheet I have, with 220K in the anode, 300V supply and 1Meg to the screen grid, this will give a stage gain of 188 times or about 7.5dB more open loop gain. Distortion is 5% at 56V rms output so if this is proportional to level, then at 30V rms output I would expect it to be about 2.7% which is close to what the 12AX7 produces. However, the extra 6dB of open loop gain means the NFB is increased by the same amount so the overall distortion should be about 6dB lower so I hope it will get below the 0.3%. Cheers Ian |
|
#36
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Alex wrote:
"Ian wrote in message ... Patrick Turner wrote: On Sep 24, 6:34 pm, wrote: "Ian wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 44x d 3.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent.http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio From the above list of performance figures we can see that there is about 1% THD per 10Vrms of output. But what is the series input resistance and to what extent would series source resistance of 100k contribute to the distortion figure of 3.6% at 36vrms output? And the answer is as follows: I used 200K in the anode and 2K2 in the cathode. HT = 313V, Plate = 170V, cathode = 1.407V With 125mV rms on the grid (2KHz) via 100K I get: 2H -59dB, 3H -77dB 4H unmeasurable The signal voltage at the plate was 10V rms which means the stage gain was 80. At the plate the measured distortion was: 2H -49dB, 3H -75dB, 4H unmeasurable So the tube itself mainly contributes 10dB of 2H The input signal was then changed to give 30V rms at the plate. Plate signal distortion was then: 2H -34dB, 3H -43dB, 4H -49dB, 5H -50dB, 6H -55dB, 7H -58dB And under the same conditions looking at the transformer secondary into 100R we get: 2H -34dB, 3H -41dB, 4H -48dB, 5H -49dB, 6H -53dB, 7H -56dB Which means the white follower plus the transformer is only contributing a couple of dB to the distortion. Also, at this point the signal voltage on the grid was just under 380mV rms and the distortion at the grid was: 2H -49, 3H -62, 4H -70, 5H -77, 6H unmeasurable which shows we are approaching grid distortion but the tube distortion is still dominant. As I mentioned before, the reason for the 100K series input resistor so I can apply shunt/shunt NFB. Connecting 2Meg from the white output back the the 12AX7 grid drops the output from 2.5V rms to 0.6V rms i.e. there's about 12dB of NFB. I increased the input to give 2.5V rms at the transformer secondary and the distortion at the transformer secondary was then: 2H-45dB, 3H -50dB, 4H -55dB, 6H -58dB, 7H -61dB which is about 11dB lower overall as expected. So, this basically seems to work. At 30V rms the 12AX7 produces about 2% distortion (-34dB) which compares well with the data sheet figure of 2.5% at 367V rms. The White follower and transformer do not seem to add significantly to this. NFB gives the expected overall reduction to just under 0.6%. I would still like the distortion to be lower so I may well try the EF86. I am not sure how much there is to be gained. From the data sheet I have, with 220K in the anode, 300V supply and 1Meg to the screen grid, this will give a stage gain of 188 times or about 7.5dB more open loop gain. Distortion is 5% at 56V rms output so if this is proportional to level, then at 30V rms output I would expect it to be about 2.7% which is close to what the 12AX7 produces. However, the extra 6dB of open loop gain means the NFB is increased by the same amount so the overall distortion should be about 6dB lower so I hope it will get below the 0.3%. Cheers Ian Hello Ian, To tackle the "mystery" of the distortion on the grid I suggested to temporarily block the plate of 12AX7 to ground by an electrolytic, say 10uF. This will reduce the gain to practically zero. After that remeasure distortion on the grid. If it reduces to nearly zero (as with the tube pulled out or cold) then it will show that the Miller effect creates (or rather magnifies) this virtual distortion. If the distortion is still high, then it is some non-linear conductance of the grid current or nonliarity of the grid capacitance due to pulsating electron cloud (space charge). Apart from a scientific curiosity you can derive something result from this experiment. For instance, it it proves Miller effect cause, then it will further inspire you to try EF86, which will not have any Miller effect at all. I would be surprised if it were a Miller effect but I will do the experiment you suggest and let you know the results. Cheers Ian If it proves to be grid current nonlinearity, then little can be done other than trying different tubes and possibly reducing the heater voltage... Alex |
|
#37
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
On Sun, 26 Sep 2010 10:36:07 +0100, Ian Bell
wrote: I would be surprised if it were a Miller effect but I will do the experiment you suggest and let you know the results. If it is Miller it will be strongly frequency dependent. That should be easy to check. d |
|
#38
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
Alex wrote: "Ian wrote in message ... Patrick Turner wrote: On Sep 24, 6:34 pm, wrote: "Ian wrote in message ... I just spent some time perusing the 12AX7 data sheets in more detail, particularly the part about RC coupled amplifiers which gives the maximum output before the onset of grid current under various conditions. Based on that data I calculated the cathode voltage and also the peak input voltage before grid current for several values of Ra, Rk and Va. Subtracting the two gives the grid bias at the onset of grid current and this seems to be pretty constant at about -0.6V. It also seems that with the 300V HT I have there is really only one operating point that will give me the 102V peak output swing I need without reaching grid current and that is with a 220K anode load, 2.2K up the cathode leg and a quiescent current of close to 0.6mA. I will try those values and see what I get. That's very close to what the "Muiderkring Electronic Tubes Handbook" (1968/p152) specifies for an ECC83 @ 300Vb and a Vo of 36Veff/102Vpp. Ra 220K R'g1 680K (load) Rk 2K2 Ia 0.63mA Vout 36Veff A 44x d 3.6% Rg 1M The same page specifies 11 other operating points for the ECC83. Can't find the book online but linked below is an equivalent.http://www.mif.pg.gda.pl/homepages/f.../054/2/288.pdf Rgds, Gio From the above list of performance figures we can see that there is about 1% THD per 10Vrms of output. But what is the series input resistance and to what extent would series source resistance of 100k contribute to the distortion figure of 3.6% at 36vrms output? And the answer is as follows: I used 200K in the anode and 2K2 in the cathode. HT = 313V, Plate = 170V, cathode = 1.407V With 125mV rms on the grid (2KHz) via 100K I get: 2H -59dB, 3H -77dB 4H unmeasurable The signal voltage at the plate was 10V rms which means the stage gain was 80. At the plate the measured distortion was: 2H -49dB, 3H -75dB, 4H unmeasurable So the tube itself mainly contributes 10dB of 2H The input signal was then changed to give 30V rms at the plate. Plate signal distortion was then: 2H -34dB, 3H -43dB, 4H -49dB, 5H -50dB, 6H -55dB, 7H -58dB And under the same conditions looking at the transformer secondary into 100R we get: 2H -34dB, 3H -41dB, 4H -48dB, 5H -49dB, 6H -53dB, 7H -56dB Which means the white follower plus the transformer is only contributing a couple of dB to the distortion. Also, at this point the signal voltage on the grid was just under 380mV rms and the distortion at the grid was: 2H -49, 3H -62, 4H -70, 5H -77, 6H unmeasurable which shows we are approaching grid distortion but the tube distortion is still dominant. As I mentioned before, the reason for the 100K series input resistor so I can apply shunt/shunt NFB. Connecting 2Meg from the white output back the the 12AX7 grid drops the output from 2.5V rms to 0.6V rms i.e. there's about 12dB of NFB. I increased the input to give 2.5V rms at the transformer secondary and the distortion at the transformer secondary was then: 2H-45dB, 3H -50dB, 4H -55dB, 6H -58dB, 7H -61dB which is about 11dB lower overall as expected. So, this basically seems to work. At 30V rms the 12AX7 produces about 2% distortion (-34dB) which compares well with the data sheet figure of 2.5% at 367V rms. The White follower and transformer do not seem to add significantly to this. NFB gives the expected overall reduction to just under 0.6%. I would still like the distortion to be lower so I may well try the EF86. I am not sure how much there is to be gained. From the data sheet I have, with 220K in the anode, 300V supply and 1Meg to the screen grid, this will give a stage gain of 188 times or about 7.5dB more open loop gain. Distortion is 5% at 56V rms output so if this is proportional to level, then at 30V rms output I would expect it to be about 2.7% which is close to what the 12AX7 produces. However, the extra 6dB of open loop gain means the NFB is increased by the same amount so the overall distortion should be about 6dB lower so I hope it will get below the 0.3%. Cheers Ian Hello Ian, To tackle the "mystery" of the distortion on the grid I suggested to temporarily block the plate of 12AX7 to ground by an electrolytic, say 10uF. This will reduce the gain to practically zero. After that remeasure distortion on the grid. If it reduces to nearly zero (as with the tube pulled out or cold) then it will show that the Miller effect creates (or rather magnifies) this virtual distortion. If the distortion is still high, then it is some non-linear conductance of the grid current or nonliarity of the grid capacitance due to pulsating electron cloud (space charge). Apart from a scientific curiosity you can derive something result from this experiment. For instance, it it proves Miller effect cause, then it will further inspire you to try EF86, which will not have any Miller effect at all. I would be surprised if it were a Miller effect but I will do the experiment you suggest and let you know the results. Cheers Ian The trouble with experiments is you get results and this time they are not as I expected, nor indeed I suspect as anyone expected. To recap with 380mV on the grid at 2KHz I got the following distortion figures at the grid: 2H -49, 3H -62, 4H -70, 5H -77 I added a 22uF 400V electrolytic from plate to ground and the figures became: 2H -60, 3H -70, 4H -77 So the distortion is reduced but by no means to zero. 2H is reduced by 11dB, 3H by 8dB and 4H by 7dB. I am not sure what the mechanism is - all I feel safe in saying is that adding a very low value ac plate load appears to reduce the distortion measured at the grid. Cheers Ian |
|
#39
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Don Pearce wrote:
On Sun, 26 Sep 2010 10:36:07 +0100, Ian wrote: I would be surprised if it were a Miller effect but I will do the experiment you suggest and let you know the results. If it is Miller it will be strongly frequency dependent. That should be easy to check. d Looks like there is some frequency dependent. I made additional tests at 100Hz, 500Hz and 4KHz as follows (grid distortion in original circuit): 100Hz 2H -63dB, 3H -65dB 500Hz 2H -58dB, 3H -61dB 2KHz 2H -49dB, 3H -62dB 4KHz 2H -46dB, 3H -65dB Looks to me like the second harmonic rises with frequency but the third varies little so I would guess the 2H is coupled somehow from the plate and the 3H is probably the onset of real grid current distortion. It might be interesting to repeat this with a grounded grid configuration. Cheers Ian |
|
#40
Posted to rec.audio.tubes
|
|||
|
|||
|
12AX7 grid current
Ian Bell wrote:
Looks like there is some frequency dependent. I made additional tests at 100Hz, 500Hz and 4KHz as follows (grid distortion in original circuit): 100Hz 2H -63dB, 3H -65dB 500Hz 2H -58dB, 3H -61dB 2KHz 2H -49dB, 3H -62dB 4KHz 2H -46dB, 3H -65dB Looks to me like the second harmonic rises with frequency but the third varies little so I would guess the 2H is coupled somehow from the plate and the 3H is probably the onset of real grid current distortion. Surely the grid current distortion is assymetric? That would make it substantially even H? It might be interesting to repeat this with a grounded grid configuration. Too many tests complicates confusion. Ian |
| Reply |
|
| Thread Tools | |
| Display Modes | |
|
|
Similar Threads
|
||||
| Thread | Forum | |||
| Screen grid mu | Vacuum Tubes | |||
| Grid Resistors | Vacuum Tubes | |||
| Grid current limit specs for "good" output tubes? | Vacuum Tubes | |||
| DC on the grid | Vacuum Tubes | |||
| Grid current in 6C33C | Vacuum Tubes | |||
![Ian Bell[_2_] is offline](images/SultanThemeVB3R/statusicon/user_offline.gif)
Linear Mode
